Motor controller

ABSTRACT

In order to propose an inexpensive and highly precise motor controller, it is structured so as to detect the position of the rotor on the basis of the difference between the real current differential vector and the reference current differential vector, thereby control the motor without using a rotation position sensor.

BACKGROUND OF THE INVENTION

The present invention relates to a motor controller.

To control the speed and torque of a synchronous motor, it is necessaryto detect or infer the pole position. By executing current control orvoltage control on the basis of the detected pole position, the speedand torque of a synchronous motor can be controlled.

In recent years, a pole position sensorless control system forcontrolling a synchronous motor without detecting the pole position ofthe synchronous motor by a position sensor is proposed.

For example, the first control method described in Japanese ApplicationPatent Laid-Open Publication No. Hei 07-245981 and Electric Society,Industry Application Department, National Convention No. 170 in the 8thyears of Heisei is a method for applying an alternating voltage andinferring the pole position on the basis of the parallel component andorthogonal component (current component in the rotatory coordinatesystem) of the motor current for the voltage and the position of themagnetic pole can be detected without using a pole position sensorduring stopping or at a low speed.

Further, the second method for superimposing an additional voltagedescribed in Japanese Application Patent Laid-Open Publication No. Hei11-150983 and Japanese Application Patent Laid-Open Publication No. Hei11-69884 is a method for realizing no-use of a pole position sensorwithin the range from low load to high load during stopping or at a lowspeed by adding an applied voltage so as to prevent magnetic saturationeven in the high torque region.

Further, the third control method described in Japanese ApplicationPatent Laid-Open Publication No. Hei 08-205578 is a method for detectingthe saliency of a synchronous motor from the mutual relation between thevector of a voltage applied to the synchronous motor by the pulse widthcontrol (PWM control) and the ripple component (current differencevector) of the motor current for it. The third method uses a general PWMsignal for controlling the voltage of the synchronous motor, so thatthere is an advantage that there is no need to load an additional signalfor detection.

Further, the voltage vector means a voltage having the magnitude anddirection decided from a three-phase voltage or d-axis and q-axisvoltages. The same may be said with the current vector and hereinafter,each phase voltage as an element or the d-axis and q-axis voltages andthe voltage vector as a sum total will be explained appropriately.Further, for the synchronous motor, the pole position of the rotor is tobe detected, so that the pole position will be explained hereunder. Fora reluctance motor, the specific position of a rotor having saliency isdetected.

Further, a control method for detecting the pole position of a rotor inthe same way as with the aforementioned method on the basis of thedifference in inductance between the q axis and the q axis using themagnetic saturation characteristic of an induction motor is proposed.

Therefore, when the aforementioned is to be described together, the poleposition and the specific position of the reluctance motor will bereferred to as a rotor position.

SUMMARY OF THE INVENTION

In the first control method mentioned above, to detect the pole positionby driving the motor, it is necessary to extract a current having thesame frequency component as that of the detection alternating voltage bya band pass filter such as a notch filter and Fourier integration.Particularly, when the number of revolutions of the motor is increased,the separation between the input frequency of the motor and thefrequency of the detection alternating voltage is difficult and aproblem arises that stable driving control at high speed rotation isdifficult. Further, it is necessary to consider so as to prevent effectby the switching characteristic of the invertor. Namely, thecarrier-frequency of the PWM signal is several kHz to 20 kHz, while thefrequency of the detection alternating voltage is low such as severalhundreds Hz, so that during driving control for the motor, noise ofseveral hundreds Hz may be generated.

Further, the second control method mentioned above is intended toimprove the characteristics for drive-controlling the motor in a stopstate or a low-speed rotation state, and the relation between thecurrent detection timing which is important for drive-controlling themotor at high-speed rotation and the PWM signal is not taken intoaccount, and highly accurate position detection is not taken intoaccount.

Further, the third control method mentioned above requires, to realizeit, to detect the mutual relation between the condition of the motorcurrent and the applied voltage every changing of the PWM signal.Namely, for one period of the carrier, it is necessary to detect themotor current condition at least 6 times and confirm the applied voltagecondition, so that a problem arises that a highly precise controllermust be used.

An object of the present invention is to propose an inexpensive andhighly precise motor controller.

Another object of the present invention is to propose a motor controllerfor controlling an AC motor with high precision by suppressing anincrease in motor loss within the wide range from stop state tohigh-speed rotation state using one current detector.

Still another object of the present invention is to propose a motorcontroller for detecting the rotor position of an AC motor withoutapplying a detection voltage to the AC motor.

The present invention has an AC motor, a power converter for applying avoltage to the AC motor by a PWM signal generated by comparing a commandvalue with a carrier, and a controller for detecting the rotor positionof the AC motor and controlling the command value and is characterizedin that the position of the rotor is detected on the basis of thedifference between the real current differential vector and thereference current differential vector.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a motor control system of the firstembodiment of the present invention.

FIGS. 2(a) and 2(b) are a time chart showing the relation between thevoltage of each phase, PWM signal, and carrier synchronizing signal whenthe detection voltage in the first embodiment shown in FIG. 1 isapplied.

FIG. 3 is a low chart of the process executed by the detectioncalculation unit in the first embodiment shown in FIG. 1.

FIG. 4 is a block diagram showing the relation between input and outputof the reference current differential calculation unit in the firstembodiment shown in FIG. 1.

FIG. 5 is a vector diagram showing the condition of the detectionvoltage vector and current differential vector in the first embodimentshown in FIG. 1.

FIG. 6 is a vector diagram showing the condition of the detectionvoltage vector and current differential vector in the first embodimentshown in FIG. 1.

FIG. 7 is a vector diagram showing the condition of the detectionvoltage vector and current differential vector in the first embodimentshown in FIG. 1.

FIG. 8 is a block diagram of a motor control system showing the secondembodiment of the present invention.

FIG. 9 is a flow chart of the process executed by the detectioncalculation unit in the second embodiment shown in FIG. 8.

FIG. 10 is a flowchart of the current sensor error detection processexecuted by the current sensor error detection unit in the secondembodiment shown in FIG. 8.

FIG. 11 is a block diagram of a motor control system showing the thirdembodiment of the present invention.

FIG. 12 is a block diagram of a motor control system showing the fourthembodiment of the present invention.

FIG. 13 is a time chart showing the relation between the voltage of eachphase, PWM signal, and carrier synchronizing signal in the fourthembodiment shown in FIG. 12.

FIG. 14 is a time chart and Lissajous waveform diagram showing therelation between the sine wave voltage of each phase and the voltagedifference vector in the fourth embodiment shown in FIG. 12.

FIG. 15 is a function block diagram of the voltage setting unit in thefourth embodiment shown in FIG. 12.

FIG. 16 is a flow chart of the process executed by the h-axis currentdifferential calculation unit in the fourth embodiment shown in FIG. 12.

FIG. 17 is a vector diagram showing the relation between the controlvoltage vector, voltage difference vector, and current differentialvector in the fourth embodiment shown in FIG. 12.

FIG. 18 is a time chart and Lissajous waveform diagram showing therelation between the sine wave voltage of each phase and the voltagedifference vector in the fourth embodiment shown in FIG. 12.

FIG. 19 is a vector diagram showing the relation between the controlvoltage vector, voltage difference vector, and current differentialvector in the fourth embodiment shown in FIG. 12.

FIG. 20 is a block diagram of a motor control system showing the fifthembodiment of the present invention.

FIG. 21 is a flow chart of the process executed by the mode decisionunit in the fifth embodiment shown in FIG. 20.

FIG. 22 is a function block diagram of the voltage setting unit in thefifth embodiment shown in FIG. 20.

FIG. 23 is a time chart and Lissajous waveform diagram showing therelation between the sine wave voltage of each phase and the voltagedifference vector in the fifth embodiment shown in FIG. 20.

FIG. 24 is a time chart and Lissajous waveform diagram showing therelation between the sine wave voltage of each phase and the voltagedifference vector in the fifth embodiment shown in FIG. 20.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The first embodiment of the present invention will be explained withreference to FIG. 1. The embodiment is structured so as to control asynchronous motor having so-called reverse saliency that the d-axisinductance Ld is smaller than the q-axis inductance Lq without using aposition sensor.

FIG. 1 is a block diagram of a motor control system for driving asynchronous motor 1 which is the first embodiment by the DC energy of abattery 2.

The DC voltage of the battery 2 is converted to a three-phase AC voltageby an inverter 3 which is a power converter and applied to thesynchronous motor 1 which is an AC motor. The applied voltage is decidedby performing the following calculation by a controller 4 which is acontroller composed of a microcomputer.

The controller 4 calculates the difference of the speed command value ωrinput from a speed command generation unit 6 from the detected motorspeed ω and performs speed control calculations at a speed controller 7on the basis of the difference. The speed controller 7 outputs controlvoltages Vuc, Vvc, and Vwc of each phase on the basis of the speedcontrol calculation results. The controller 4 adds detection voltagesVus, Vvs, and Vws of each phase, which will be described later,respectively to these control voltages Vuc, Vvc, and Vwc, generatesvoltage command values Vur, Vvr, and Vwr of each phase, and inputs themto a PWM signal generation unit 8.

The PWM signal generation unit 8 generates PWM signals Pu, Pv, and Pw ofeach phase corresponding to the voltage command values Vur, Vvr, and Vwrof each phase and supplies them to the inverter 3 and the inverter 3generates output voltages corresponding to the PWM signals Pu, Pv, andPw and applies them to the synchronous motor 1.

FIG. 2 shows the relation between the voltage command values Vur, Vvr,and Vwr of each phase and the PWM signals Pu, Pv, and Pw. The PWM signalgeneration unit 8 compares the carrier of triangular waveform and thevoltage command values Vur, Vvr, and Vwr, thereby generates the PWMsignals Pu, Pv, and Pw.

The PWM signal generation unit 8 internally fetches and sets the voltagecommand values Vur, Vvr, and Vwr of each phase at the point of time(times t1, t3, t5, . . . ) when the carrier takes the maximum value,compares them with the carrier, and generates the PWM signals Pu, Pv,and Pw.

The waveform when the voltage command values Vur, Vvr and Vwr (=thecontrol voltages Vuc, Vvc, and Vwc) to which the detection voltages Vus,Vvs, and Vws are not added are fetched is as shown in FIG. 2(a).

On the other hand, when the detection voltages Vus, Vvs, and Vws areadded (applied to the synchronous motor 1), the PWM signal generationunit 8 sets the positive and negative detection voltages Vus, Vvs, andVws every a half period (times t1, t2, t3, . . . ) of the carrier so asto obtain the waveform shown in FIG. 2(b). Namely, in the section 1shown in FIG. 2(b), the PWM signal generation unit 8 adds (applies) thedetection voltages Vus, Vvs, and Vws so as to set the detection voltagevector in the detection voltage direction θv which will be describedlater. Further, in the section 2 shown in FIG. 2(b), the PWM signalgeneration unit 8 adds the detection voltages Vus, Vvs, and Vws so as toapply the detection voltage vector in the opposite direction (in thedirection different by 180 degrees) to the detection voltage direction.

FIG. 3 is a flow chart showing the process to be executed by a detectionvoltage calculation unit 10 to realize it.

Step 101

The detection voltage direction θv is obtained by calculation of 2θc/2.The reason for the voltage direction will be described later byreferring to FIGS. 5 to 7.

Step 102

The voltage application timing is judged and the process is branched.Namely, as shown in FIG. 2, when the times t1, t3, . . . are judged as apoint of time when the carrier reaches the maximum value, the process isbranched to Step 103 and when the times t2, t4, . . . are judged as apoint of time when the carrier reaches the minimum value, the process isbranched to Step 104.

Step 103

To set the vector Vs of detection voltage applied to the synchronousmotor 1 in the section 1 in the detection voltage direction θv (positivedirection), the detection voltages Vus, Vvs, and Vws of each phase arecalculated.

Step 104

To set the vector Vs of detection voltage applied to the synchronousmotor 1 in the section 2 in the detection voltage direction θv (negativedirection, that is, direction of θv+π), the detection voltages Vus, Vvs,and Vws of each phase are calculated.

In FIG. 3, Vs0 for deciding the magnitude of the detection voltagevector Vs is set to ½, thus the voltage of each phase is decided. Thereason is that the voltage difference between the sections 1 and 2 isdefined a real detection voltage vector Vs. Further, it is desirable toset Vs0 to a small value as far as possible as long as the variation ofcurrent can be detected. Further, in this case, on the basis of the αaxis of the α−β static coordinate system having the orthogonal α axisands axis, the phase or direction is decided and the U phase is set onthe α axis. Therefore, the directions of the V and W phases aredirections of 2π/3 and 4π/3 to the α axis respectively.

Next, the detection method for the rotor position in the firstembodiment shown in FIG. 1 will be explained.

As a current sensor 5 u for detecting the U-phase current of thesynchronous motor 1, an inexpensive current transformer CT for detectingonly the AC component of a current flowing in the U phase is used. Bydoing this, only the pulsating component of a current by the PWM signalis detected.

A current detection unit 9 of the controller 4 fetches and detects aU-phase current iu output from the current transformer CT in the timingcoinciding with a carrier synchronous signal P1 synchronized with themaximum value and minimum value of the carrier.

A current differential calculation unit 11 obtains the variation of theU-phase current iu from the detection voltage vector, that is, theU-phase current difference Δiu as shown below. The current differentialcalculation unit 11 calculates the current difference Δiu1 in thesection 1 shown in FIG. 2 from the difference between the U-phasecurrent iu1 fetched by the current detection unit 9 at the point of timeof the maximum value of the carrier (for example, time t1) and theU-phase current iu2 fetched at the point of time of the next minimumvalue of the carrier (time t2). Further, the current differentialcalculation unit 11 calculates the current difference Δiu2 in thesection 2 shown in FIG. 2 from the difference between the U-phasecurrent iu2 and the U-phase current iu3 fetched at the point of time ofthe next maximum value of the carrier (time t3). The current differencesΔiu1 and Δiu2 are affected by the control voltages Vuc, Vvc, and Vwc,the detection voltage vector, and the counter electromotive force of thesynchronous motor 1. However, in consideration of the difference betweenthe current differences Δiu1 and Δiu2, when the applied voltage andcounter electromotive force are the same, their effects are canceled.

Therefore, as explained by referring to FIG. 2, in the sections 1 and 2,when the control voltages Vuc, Vvc, and Vwc are applied in the samevalue and only the detection voltage vectors Vs are applied in differentvalues, the U-phase current difference Δiu which is the differencebetween the current differences Δiu1 and Δiu2 is affected only by thedifference in the detection voltage vector Vs between the sections 1 and2. Namely, the variation of the U-phase current iu to the detectionvoltage vector Vs and the U-phase current difference Δiu can be detectedquite independently of the control voltages Vuc, Vvc, and Vwc.Hereinafter, the difference in the detection voltage vector between thesections 1 and 2 is called a detection voltage vector Vs.

Meanwhile, when the rotor position θ and the d-axis and q-axisinductances of the synchronous motor 1 are known, the variation of theU-phase current iu to the detection voltage vector Vs can be obtained bycalculation. This value is assumed as a U-phase reference currentdifference Δicu. Actually, instead of the rotor position θ, the inferredrotor position θc calculated by the controller 4 is known, so thatassuming that the inferred rotor position θc agrees with the rotorposition θ, the U-phase reference current difference Δicu is obtained bya reference current differential calculation unit 12. In this process,as shown in FIG. 4, when a table is prepared for the inferred rotorposition θc, the U-phase reference current difference Δicu can beobtained simply. This obtaining method will be described later togetherwith the vector diagrams of FIGS. 5 to 7.

The difference between the detected U-phase current difference Δiu andthe U-phase reference current difference Δicu (hereinafter, this iscalled a U-phase detection current difference Δisu) indicates avariation (difference) between the inferred rotor position θc and therotor position θ, so that a position detection unit 13 controlsconverging by using a control means such as proportion-integrationcalculations so as to set the difference to 0.

It is an important point of the present invention to make the relationthereof clear and it will be described later by referring to the vectordiagrams of FIGS. 5 to 7.

The inferred rotor position θc obtained as mentioned above is input to aspeed detection unit 14 and used to obtain the motor speed from thevariation thereof. Further, the inferred rotor position θc is input tothe speed control unit 7 and also used to output the control voltagevector obtained by the speed control unit 7 to the control voltages Vuc,Vvc, and Vwc of each phase by coordinate conversion.

Next, the detection of the rotor position θ in the motor control systemshown in FIG. 1 will be explained by referring to FIG. 5.

FIG. 5 shows a state that the d−q axis rotatory coordinate system thatthe pole position is on the d axis rotates from the α axis by the rotorposition θ and the inferred rotor position θc of the controller 4 islarger than the real rotor position θ and different from the real rotorposition θ. The ellipse indicated by a solid line that the d axis is along axis and the q axis is a short axis indicates a Lissajous waveformof the current differential vector Δi to the detection voltage vector Vswhen the detection voltage vector Vs makes one revolution from 0 to 2π.Therefore, when the inductances of the d axis and q axis of thesynchronous motor 1 are the set values, the actually-generated currentdifferential vector Δi to the detection voltage vector Vs moves on thesolid-line ellipse. On the other hand, when the inductances of the daxis and q axis of the synchronous motor 1 are as set and the rotorposition of the synchronous motor 1 coincides with the inferred rotorposition θc inferred by the controller 4, the current differentialvector Δi moves on the ellipse indicated by a dashed line that the dcaxis is a long axis and the qc axis is a short axis. This is called areference current differential vector Δic.

In this state, as shown in FIG. 5, the change condition of current whenthe detection voltage vector Vs is applied in the detection voltagedirection θv, that is, in the direction of 2θc+π/2 will be explained.

When the detection voltage vector Vs is applied in the phase θvdirection, the current differential vector Δi actually generated, asshown in FIG. 5, is a vector on the Lissajous waveform line indicated bya solid line and expressed by the following formula.Δi=Δiα+jΔiβ  (Formula 1)

where j means an imaginary axis and Δiα and Δis mean the followingformulas.Δiα=Δids·cos θ−Δiqs·sin θ  (Formula 2) Δis=Δids·sin θ+Δiqs·cos θ  (Formula 3)Δids=Vs0·cos(θv−θ)Δt/Ld  (Formula 4)Δiqs=Vs0·sin(θv−θ)Δt/Ld  (Formula 5)where Ld and Lq indicate the inductances of the d axis and q axis of thesynchronous motor 1 respectively and Vs0 indicates the magnitude of thedetection voltage (length of the detection voltage vector Vs shown inFIG. 5). Therefore, Formula 2 and Formula 3 are expressed as follows.Δiα=(½)·Vs0·Δt{(1/Ld+1/Lq)cos θv+(1/Ld−1/Lq)cos(θv−2θ)}  (Formula 6)Δis=(½)·Vs0·Δt{(1/Ld+1/Lq)sin θv−(1/Ld−1/Lq)sin(θv−2θ)}  (Formula 7)

In the same way, when the real rotor position coincides with theinferred rotor position θc inferred by the controller 4, the referencecurrent differential vector Δic generated by applying the detectionvoltage vector Vs in the phase θv direction is on the Lissajous waveformindicated by a dashed line. In FIG. 5, the detection voltage vector Vsis closer to the qc axis than the q axis, so that the reference currentdifferential vector Δic is a vector closer to the detection voltagevector Vs than the current differential vector Δi and expressed by thefollowing formula.Δic=Δiαc+jΔiβc  (Formula 8)

where Δiαc and Δiβc mean the following formulas.Δiαc=Δidsc·cos θc−Δiqsc.sin θc  (Formula 9) Δiβc=Δidsc·sin θc+Δiqsc.cos θc  (Formula 10)Δidsc=Vs0·cos(θv−θc)Δt/Ldc  (Formula 11)Δiqsc=Vs0·sin(θv−θc)Δt/Ldc  (Formula 12)

where Ldc and Lqc indicate the inductances of the reference d axis and qaxis of the synchronous motor 1 set by the controller 4. Therefore,Formula 9 and Formula 10 are expressed as follows.Δiαc=(½)·Vs0·Δt{(1/Ldc+1/Lqc)cos θv+(1/Ldc−1/Lqc)cos(θv−2θc)}  (Formula13)Δisc=(½)·Vs0·Δt{(1/Ldc+1/Lqc)sin θv−(1/Ldc−1/Lqc)sin(θv−2θc)}  (Formula14)

In this case, the U-phase reference current difference Δicu shown inFIG. 4 is the U-phase contribution of Δiαc and a value proportional toΔiαc, so that a table can be prepared by calculation including thedetection voltage direction θv. Further, the detection voltage directionθv is set to a value of (2θc+π/2), though it will be described later.Therefore, Formula 3 is expressed as follows, so that a table forobtaining the U-phase reference current difference Δicu is prepared onthe basis of Formula 15.Δiαc=(½)·Vs0·Δt(1/Ldc+1/Lqc)cos(2θc+π/2)  (Formula 15)

Next, the difference between the current differential vector Δi and thereference current differential vector Δic and the detection currentdifferential vector Δis will be examined. Further, the inductances Ldcand Lqc of the reference d axis and q axis of the synchronous motor 1are respectively different from the inductances Ld and Lq of the real daxis and q axis and a method taking it into account may be usedtogether. However, in this case, Ldc=Ld and Lqc=Lq are assumed to beheld.

The following may be obtained using Formula 1 to Formula 4.Δis=Δi−Δic≡Δiαs+jΔiss  (Formula 16)

where Δiαs and Δiss are given the following formulas.Δiαs=(½)·Vs0·Δt(1/Ld−1/Lq){cos(θv−2θ)−cos(θv−2θc)}  (Formula 17)Δisc=(½)·Vs0·Δt(1/Ld−1/Lq){sin(θv−2θ)−sin(θv−2θc)}  (Formula 18)

Then, when (2θc+π/2) is substituted for the detection voltage directionθv, Formula 17 and Formula 18 can be expanded as shown below. Further,(½).Vs0.Δt(1/Ld−1/Lq) is assumed as a constant of K0. $\begin{matrix}\begin{matrix}{{\Delta\quad i\quad\alpha\quad s} = {{K0}\left\{ {{\cos\left( {{\theta\quad v} - {2\theta}} \right)} - {\cos\left( {{\theta\quad v} - {2\theta\quad c}} \right)}} \right\}}} \\{= {{K0}\left\{ {{\cos\left( {{2\theta\quad c} - {2\theta} + {\pi/2}} \right)} - {\cos\left( {\pi/2} \right)}} \right\}}} \\{= {{{- {K0}} \cdot \sin}\quad 2\left( {{\theta\quad c} - \theta} \right)}} \\{= {{{- {K0}} \cdot \sin}\quad 2\left( {\theta - {\theta\quad c}} \right)}}\end{matrix} & \left( {{Formula}\quad 19} \right) \\\begin{matrix}{{\Delta\quad i\quad\beta\quad s} = {{- {K0}}\left\{ {{\sin\left( {{\theta\quad v} - {2\theta}} \right)} - {\sin\left( {{\theta\quad v} - {2\theta\quad c}} \right)}} \right\}}} \\{= {{- {K0}}\quad{K0}\left\{ {{\sin\left( {{2\theta\quad c} - {2\theta} + {\pi/2}} \right)} - {\sin\left( {\pi/2} \right)}} \right\}}} \\{= {{K0} \cdot \left\{ {1 - {\cos\quad 2\left( {{\theta\quad c} - \theta} \right)}} \right\}}}\end{matrix} & \left( {{Formula}\quad 20} \right)\end{matrix}$

The vector expressed by Formula 19 and Formula 20 is the detectioncurrent differential vector Δis. Therefore, as shown in FIG. 5, when theinferred rotor position θc is larger than the real rotor position θ, ifthe detection voltage vector Vs is applied assuming that the detectionvoltage direction θv is (2θc+π/2), the detection current differentialvector Δis is a vector in the direction close to the negative directionof the α axis. Particularly, in consideration of Formula 19, when thedetection voltage vector Vs is applied assuming that the detectionvoltage direction θv is (2θc+π/2), the α axis component Δiαs of thedetection current differential vector Δis is a value proportional to sin2(θ−θc). Therefore, when the α axis component Δiαs is set to 0, theinferred rotor position θc can coincide with the real rotor position θ.Further, in this embodiment, the U phase of the synchronous motor 1coincides with the α axis, the α axis component Δiαs of the detectioncurrent differential vector Δis is proportional to the U-phase detectioncurrent differential vector Δisu. Therefore, in FIG. 5, the U-phasedetection current differential vector Δisu is negative and it means thatthe inferred rotor position θc is larger than the real rotor position θ,so that by performing control calculations so as to reduce the inferredrotor position θc, the inferred rotor position θc can be brought closeto the real rotor position θ. Such calculations are performed by theposition detection unit 13.

FIG. 6 shows the relation between the detection voltage vector Vs andthe current differential vectors Δi, Δic, and Δis for it when theinferred rotor position θc is close to the real rotor position θ in thisway. Since the inferred rotor position θc is made smaller, it is foundthat the direction θv of the detection voltage vector Vs is smaller thanthat shown in FIG. 5. Therefore, the current differential vector Δi andthe reference current differential vector Δic respectively move in thedifferent directions from those shown in FIG. 5, while the detectioncurrent differential vector Δis is directed almost in the negativedirection of the α axis and the magnitude thereof is made smaller.Therefore, it is found that the inferred rotor position θc is close tothe real rotor position θ. On the basis of the U-phase detection currentdifferential vector Δisu proportional to the α axis component thereof,the inferred rotor position θc can coincide with the real rotor positionθ by control calculation.

When the inferred rotor position θc is small for the real rotor positionθ as shown in FIG. 7, the detection current differential vector Δis isdirected close to the positive direction of the α axis, so that theinferred rotor position θc can coincide with the real rotor position θby control calculation in the same way. These relations can be derivedfrom Formula 19 and Formula 20.

In this embodiment, the rotor position can be inferred precisely usingone inexpensive current sensor, so that compared with a conventionalcontroller using a plurality of current sensors, an inexpensive positionsensorless controller can be realized. Moreover, the inference can beexecuted by a calculation process on the basis of a one-phase current,so that the controller 4 can be realized using an inexpensivemicroprocessor.

Further, the current sensor 5 u may detect only the pulsating componentof a current on the basis of the PWM signal, so that a signal having afrequency component in the neighborhood of the carrier frequency isinput to the current detection unit 9, thus the resolution of currentdetection can be improved. By doing this, the magnitude of the detectionvoltage vector Vs can be reduced and there is an advantage that theeffect by addition of the detection voltage can be reduced greatly.

FIG. 8 is a block diagram of a motor control system showing the secondembodiment of the present invention. In this embodiment, as comparedwith the first embodiment, the method for applying the detection voltagevector Vs is different in a point that two current sensors 5 v and 5 ware used for current control for a torque command instead of a speedcommand. The method for inferring the rotor position using one currentsensor 5 u is the same as that of the first embodiment, so thatduplicate explanation will be omitted.

This embodiment is suitable for motor control of an electric car forgenerating torque proportional to a torque command τr according to thestepping depth of an accelerator pedal.

When the torque command τr is input to the current command unit 16, thecurrent command unit 16 calculates ad-axis current command value idr forcontrolling the magnetic flux of the synchronous motor 1 and a q-axiscurrent command value idq orthogonal to it on the basis of the torquecommand Tr and the motor speed ω. The calculation may be performed so asto obtain from a table prepared by calculating the d-axis currentcommand value idr and the q-axis current command value idq orthogonal toit beforehand so as to minimize the loss of the driving system of thesynchronous motor 1 for the torque command τr and the motor speed ω. Thed-axis current command value idr and the q-axis current command valueidq obtained here are input to the current control unit 17.

Further, a V-phase current iv and a W-phase current iw which aredetected by the current sensors 5 v and 5 w are converted from analogueto digital by the current detection unit 15 and fetched inside thecontroller 4 as a digital amount. Thereafter, by the coordinateconversion unit 19, these currents are coordinate-converted from thestatic coordinate system to the d−q axis rotatory coordinate systemrotating in the same way as with the rotor using the inferred rotorposition θc obtained by the position detection unit 13 and a d-axiscurrent id and a q-axis current id are obtained.

The d-axis current id and the q-axis current id are input to the currentcontrol unit 17, and the feedback control calculation by the differencebetween the d-axis current command value idr and the d-axis current idis performed by the current control unit 17, thus a d-axis controlvoltage Vdc is decided, and the feedback control calculation by thedifference between the q-axis current command value iqr and the q-axiscurrent iq is performed, thus a q-axis control voltage Vqc is decided.The control calculation is generally the proportion-integrationcalculation. Further, as a method for correcting the counterelectromotive force accompanying rotation of the synchronous motor 1,non-interference control according to the motor speed ω may be usedtogether.

From the viewpoint of control, when the detection voltages Vqs and Vdsare ignored, the d-axis voltage command value Vdr (=d-axis controlvoltage Vdc) and the q-axis voltage command value Vqr (=q-axis controlvoltage Vqdc) are converted from the d−q axis rotatory coordinate systemto the α−β axis static coordinate system by the coordinate conversionunit 18 and 3-phase voltage commands Vur, Vvr, and Vwr are output. Bythis addition of the current control system, the d-axis current id cancoincide with the d-axis current command value idr and the q-axiscurrent id can coincide with the d-axis current command value idrrespectively at a rapid response speed.

In the second embodiment, the position sensorless control system canrealize rapid-response torque control.

Further, the detection voltage calculation unit 20 in the secondembodiment is structured so as to apply the detection voltages Vds andVqs in the d−q axis coordinate system. The processing method executed bythe detection voltage calculation unit 20 will be explained by referringto FIG. 9.

FIG. 9 is a flow chart of the process executed by the detection voltagecalculation unit 20.

Step 111

The detection voltage direction θv is obtained by calculation of(θc+π/2). The reason is that a detection voltage is applied to the d−qaxis rotatory coordinate system and as shown in the vector diagrams inFIGS. 5 to 7, the phase difference between the a−s axis staticcoordinate system which is a static coordinate system and the dc−qc axisrotatory coordinate system is θc.

Step 112

In the same way as with Step 102 mentioned above, the detection voltagecalculation unit 20 judges the timing of voltage application andbranches the process. In this case, the detection voltage is changedevery a half period of the carrier. However, when the application periodof the control voltage is equal to two periods of the carrier, thejudgment change at Step 112 may be executed every one period of thecarrier.

Step 113

The detection voltages Vds and Vqs applied at the point of time when thecarrier is maximized are calculated.

Step 114

The detection voltages Vds and Vqs applied at the point of time when thecarrier is minimized are calculated.

The processing method shown in FIG. 9 has an advantage compared with theprocessing method shown in FIG. 3 that there are very few calculationcontents.

Further, the second embodiment is added with a current sensor errordetection unit 21. Generally, a method for detecting an error of thecurrent sensor using that the sum of three-phase currents is 0 is known.However, in the second embodiment shown in FIG. 8, in the same way aswith the first embodiment shown in FIG. 1, the current sensor 5 u ofU-phase is structured so as to use an inexpensive sensor having afunction for detecting only the AC amount, so that to detect existenceor no-existence of an error of the current sensor, a new detectionmethod must be designed.

FIG. 10 is a flowchart of the current sensor error detection processexecuted by the current sensor error detection unit 21.

Step 121

The U-phase current difference Δiu is obtained from the differencebetween the U-phase current iu(n) at the time t(n) and the U-phasecurrent iu(n−1) at the time t(n−1). In this case, the time (n) means thepoint of time of the minimum value of the carrier and concretely, it isequivalent to the times t2 and t4 shown in FIG. 2. Further, the time(n−1) means the point of time of the maximum value of the carrier and inthe same way, it is equivalent to the times t1 and t3 shown in FIG. 2.Further, with respect to the U-phase current, the current sensor 5 u fordetecting only the AC component is used, so that a value different fromthe current actually flowing is detected, while the U-phase currentdifference Δiu which is a fluctuation component is the same as the realvalue.

Step 122

The U-phase current difference Δiv is obtained in the same way.

Step 123

The W-phase current difference Δiw is obtained in the same way.

With respect to the V phase and W phase, for the purpose of execution ofcurrent control during stop and at a low speed, a current sensor capableof also detecting the DC component may be used.

Step 124

The sum total Δi0 of 3-phase current differences is calculated. In ageneral case that a zero-phase current does not flow in the synchronousmotor 1, the sum total of 3-phase currents is 0, so that the sum totalΔi0 of 3-phase current differences is also 0.

Step 125

Whether the sum total Δi0 of current differences is less than apredetermined decided value Δij or not is decided. When the sum totalΔi0 is less than the decided value Δij, the current sensor errordetection unit 21 judges that the current sensor is normal and ends theprocess, and when the sum total Δi0 is the decided value Δij or more,the process is branched to Step 126.

Step 126

The current sensor error detection unit 21 generates a current sensorerror signal Sc and inputs it to the PWM signal generation unit 8. ThePWM signal generation unit 8, when the current sensor error signal Sc isinput, stops generation of the PWM signals Pu, Pv, and Pw and stops thesynchronous motor 1.

When the current sensors 5 u, 5 v, and 5W enter an error state likethis, the synchronous motor 1 is stopped, thus the high reliability ofthe position sensorless control system for executing highly efficientcurrent control is ensured.

FIG. 11 is a block diagram of a motor control system showing the thirdembodiment of the present invention. This embodiment is an embodimentthat a motor control system having the equivalent performance to that ofthe motor control system of the second embodiment is structured at a lowprice. For the constitution duplicated with the aforementionedembodiment, the explanation will be omitted.

The third embodiment has a constitution that one current sensor 5 x fordetecting the input current of the inverter 3 is used instead of thecurrent sensors 5 v and 5 w, and the input current iDC detected by thecurrent sensor 5 x and the PWM signals Pu, Pv, and Pw of each phase areinput to a phase current separation unit 22, and the V-phase current ivand the W-phase current iw are obtained by calculation.

From the logic of the PWM signals Pu, Pv, and Pw of three-phase, therelation between the input current iDC and the currents of each phase isfound. For example, when the PWM signal Pu is on the high level and thePWM signals Pv and Pw are on the low level, the power element on theupper side of the V phase and the power elements on the lower side ofthe U phase and W phase in the 3-phase bridge circuit of the inverter 3are turned on, so that the input current iDC agrees with the positiveV-phase current. Further, when the PWM signals Pu and Pv are on the highlevel and the PWM signal Pw is on the low level, the power elements onthe upper side of the U phase and V phase and the power element on thelower side of the W phase are turned on, so that the input current iDCagrees with the negative W-phase current. By tabling the relationbetween this pattern of the PWM signals Pu, Pv, and Pw and the phasecurrents, the current of each phase can be obtained on the basis of thedetected input current iDC and the PWM signals Pu, Pv, and Pw. The phasecurrent separation unit 22 obtains the V-phase current iv and theW-phase current iw on the basis of this relation between the inputcurrent iDC, the PWM signals Pu, Pv, and Pw, and each phase current.This constitution can reduce the number of current sensors to be used.Further, the current sensor 5 u for position detection is used to detecta current of one phase (U phase in this case) every predetermined timingsynchronized with the carrier.

According to the controller of the third embodiment, a rapid-responseposition sensorless control system can be realized at a low price.

FIG. 12 is a block diagram of a motor control system showing the fourthembodiment of the present invention. This embodiment is structured so asto detect the phase current without applying a detection voltage at thetiming showing the same phenomenon as that when the detection voltage isapplied, thereby realize detection of the rotor position using thesaliency of the synchronous motor 1, prevent an increase in the loss dueto an increase in noise and pulsation of the current by applying thedetection voltage, and realize a rapid-response position sensorlesscontrol system at a low price.

Therefore, the current differential vector is detected using two currentsensors. The fourth embodiment shown in FIG. 12 constitutes a currentcontrol system for the torque command τr in the same way as with thesecond embodiment shown in FIG. 8. However, a method for applying nodetection voltage, obtaining a voltage vector instead of it, andinferring the rotor position θc is different. For the complicateconstitution with the aforementioned embodiment, the explanation will beomitted.

In the fourth embodiment, the PWM signal generation unit 8 generates acarrier synchronizing signal P2 for current detection at timingdifferent from that of the carrier synchronizing signal P1 of theaforementioned embodiment. The carrier synchronizing signal P2, as shownin the time chart in FIG. 13, is generated so as to detect the currentof each phase at the times ta, tb, tc, and td when the carrier takes anintermediate value.

The current detection unit 15 fetches the V-phase current iv and theW-phase current iw at the generation timing of the carrier synchronizingsignal P2. The application condition of the voltage of each phase atthis time will be explained by referring to FIG. 13. Further, during theperiod from the time t1 to t5, it is assumed that the voltages (controlvoltages Vur, Vvr, and Vwr) to be applied to each phase are not changed.

The mean voltage of each phase during the period from the time t1 to t5(or t3) is naturally Vur, Vvr, and Vwr respectively. However, as the PWMsignals Pu, Pv, and Pw, show, in the section A between the times ta andtb, the U-phase voltage and V-phase voltage are maximum values and theW-phase voltage is a negative value, while in the section B between thetimes tb and tc, the U-phase voltage is close to 0, and the V-phasevoltage is a negative value, and the W-phase voltage is a minimum value.Namely, between the section A and the section B, there is a differencein the voltage vector to be applied. This is called a voltage differencevector ΔVs. In the time chart of the first embodiment shown in FIG. 2,the positive and negative detection voltage vectors are additionallyapplied in the sections 1 and 2 so as to generate a difference in thevoltage vector. However, when the current detection timing is set as inthe fourth embodiment, an equivalent state to that when the detectionvoltage is applied can be set.

Next, the relation between the control voltage vector Vc and the voltagedifference vector ΔVs will be explained by referring to FIG. 14. FIG.14(a) shows waveforms when the voltage of each phase for the phase ofthe control voltage vector is a sine waveform. In this case, theLissajous waveforms of the voltage vectors in the sections A and B arerespectively as shown in (b) and (c) and formed like a bulged triangle.In this case, the arrows shown in (b) and (c) indicate voltage vectorswhen the phase of the control voltage vector is 150 degrees. Further,the mean voltage vector in the sections A and B is a circle as shown in(d) and the phase of the mean voltage vector indicated by the arrow is150 degrees. The average of the two voltage vectors indicated by thearrows in (b) and (c) is the mean voltage vector in (d). The Lissajouswaveform of the mean voltage vector is naturally circular because itindicates a voltage vector of a sine wave.

The Lissajous waveform of the voltage difference vector ΔVs is as shownin (e) and the voltage difference vector ΔVs when the phase of thecontrol voltage vector is 150 degrees is directed in the direction of−120 degrees. When the phase of the control voltage vector is set as atransverse axis, and the absolute value ΔVs0 of the voltage differencevector ΔVs and the phase Δv thereof are set as an ordinate axis, and theLissajous waveform of the voltage difference vector ΔVs shown in (e) isconverted, it is as shown in (f). The absolute value ΔVs0 of the voltagedifference vector ΔVs pulsates in a period of ⅙ times of one period ofthe phase of the control voltage vector and the phase θv of the voltagedifference vector ΔVs rotates two times in one period of the phase ofthe control voltage vector. When the control voltages Vur, Vvr, and Vwrof each phase are decided, the voltage difference vector ΔVs is decideduniquely and the control voltage vector can be calculated by tabling thephase thereof.

Therefore, in the fourth embodiment, when the control voltages Vd and Vqof the d and q axes decided by the current control unit 17 and theinferred rotor position θc output from the position detection unit 13are input to the voltage setting unit 25, the voltage command valuesVur, Vvr, and Vwr of each phase are calculated and the absolute valueΔVs0 of the voltage difference vector ΔVs and the phase θv thereof canbe obtained by the table. These values are output to perform positiondetection calculations.

The voltage setting unit 25 will be explained by referring to thefunction block diagram shown in FIG. 15. The voltage vector calculationunit 27 obtains the absolute value Vc0 of the control voltage vector andthe vector phase δ from the dc axis on the basis of the control voltagesVd and Vq. By adding the vector phase θ and the inferred rotor positionθc, the phase θvc from the α axis of the control voltage vector isobtained, and then the single-phase voltage table 28 is referred to, andthe single-phase voltages vu, vv, and vw which are the basis for theapplied voltages of each phase are obtained. In this case, the waveformsof sine wave shown in FIG. 14(a) are tabled.

Further, the absolute value ΔVs0 of the voltage difference vector ΔVsand the phase θv thereof, as shown in FIG. 14(f), are decided by theabsolute value Vc0 of the control voltage vector and the phase θvcthereof. Then, in the single-phase voltage table 28, the phase θv of thevoltage difference vector for the phase θvc and the unit voltagedifference vs which is the absolute value ΔVs0 of the voltage differencevector when the absolute value Vc0 of the control voltage vector is 1 Vare tabled, thus the unit voltage difference vs and the phase θv arecalculated. The magnitude of voltage is proportional to the absolutevalue Vc0 of the control voltage vector, so that the products of theabsolute value Vc0 of the control voltage vector, the single-phasevoltages vu, vv, and vw, and the unit voltage difference vs arecalculated respectively by the multiplication unit 29 and the voltagecommand values Vur, Vvr, and Vwr of each phase and the absolute valueΔVs0 of the voltage difference vector are obtained.

Next, a method for inferring the rotor position from the voltagedifference vector will be explained. In the first embodiment shown inFIG. 1, there is a degree of freedom for applying the detection voltagevector Vs in an optional direction and it is used. Namely, the directionof the detection voltage vector Vs is decided so that a detectioncurrent difference Δis for an error of the rotor position appears in aspecific direction (in the first embodiment shown in FIG. 1, thedirection of the α axis). However, the voltage difference vector for thedetection voltage vector Vs is uniquely decided when the control voltagevector is decided, so that it cannot be set in an optional direction.Therefore, the fourth embodiment has a constitution that inversely forthe decided voltage difference vector and the inferred rotor positionθc, the direction that the detection current difference Δis for an errorof the rotor position appears is identified and control for setting thecomponent of the detection current difference Δis in the direction to 0is executed. Further, the direction that the detection currentdifference Δis appears is set as an h axis here and the phase thereof isset to θh.

When the calculation method for Formula 1 to Formula 20 is used, thephase θh of the h axis is obtained by the following formula.θh=2θc−θv+π/2  (Formula 21)

The h-axis phase calculation unit 26 in the fourth embodiment calculatesFormula 21 on the basis of the inferred rotor position θc and thevoltage difference vector phase θv and outputs the phase θv of the haxis. In this case, the h-axis reference current difference Δihc whichis the θh component in the h-axis direction of the reference currentdifferential vector Δic is obtained by the following formula.$\begin{matrix}\begin{matrix}{{\Delta\quad{ic}} = {{\Delta\quad i\quad\alpha\quad{c \cdot \cos}\quad\theta\quad h} + {\Delta\quad i\quad\beta\quad{c \cdot \sin}\quad\theta\quad h}}} \\{= {{\left( {1/2} \right) \cdot {Vs0} \cdot \Delta}\quad{t \cdot \left\lbrack \left\{ {{\left( {{1/{Ldc}} + {1/{Lqc}}} \right)\cos\quad\theta\quad v} +} \right. \right.}}} \\{{{\left. {\left( {{1/{Ldc}} + {1/{Lqc}}} \right){\cos\left( {{\theta\quad v} - {2\theta\quad c}} \right)}} \right\}}\cos\quad\theta\quad h} +} \\{\left\{ {{\left( {{1/{Ldc}} + {1/{Lqc}}} \right)\sin\quad\theta\quad v} - \left( {{1/{Ldc}} - {1/{Lqc}}} \right)} \right.} \\\left. {\left. {\left. {\sin\left( {{\theta\quad v} - {2\theta\quad c}} \right)} \right)} \right\}\sin\quad\theta\quad h} \right\rbrack \\{= {{\left( {1/2} \right) \cdot {Vs0} \cdot \Delta}\quad{t \cdot \left\lbrack {\left( {{1/{Ldc}} + {1/{Lqc}}} \right)\cos\quad\theta\quad v} \right.}}} \\{{\left. {{\cos\quad\theta\quad h} + {\sin\quad\theta\quad v\quad\sin\quad\theta\quad h}} \right)} + \left( {{1/{Ldc}} - {1/{Lqc}}} \right)} \\{\left. \left\{ {{\cos\quad\left( {{\theta\quad v} - {2\theta\quad c}} \right)\cos\quad\theta\quad h} - {{\sin\left( {{\theta\quad v} - {2\theta\quad c}} \right)}\sin\quad\theta\quad h}} \right\} \right\rbrack} \\{= {{\left( {1/2} \right) \cdot {Vs0} \cdot \Delta}\quad{t \cdot \left\lbrack {\left( {{1/{Ldc}} + {1/{Lqc}}} \right)\cos} \right.}}} \\{\left. {\left( {{\theta\quad v} - {\theta\quad h}} \right) + {\left( {{1/{Ldc}} - {1/{Lqc}}} \right){\cos\left( {{\theta\quad v} - {2\theta\quad c} + {\theta\quad h}} \right)}}} \right\rbrack} \\{= {{\left( {1/2} \right) \cdot {Vs0} \cdot \Delta}\quad{t \cdot \left( {{1/{Ldc}} + {1/{Lqc}}} \right)}\cos}} \\{\left( {{2\theta\quad v} - {2\theta\quad h} - {\pi/2}} \right)} \\{= {{\left( {1/2} \right) \cdot {Vs0} \cdot \Delta}\quad{t \cdot \left( {{1/{Ldc}} + {1/{Lqc}}} \right)}\sin\quad 2\left( {{\theta\quad v} - {\theta\quad c}} \right)}}\end{matrix} & {{Formula}\quad 22}\end{matrix}$

where Vs0 is the absolute value of the voltage differential vector ΔVs.The absolute value ΔVs0, as shown in FIG. 14(f), varies with the phaseof the control voltage vector, so that the h-axis reference currentdifferential calculation unit 23 obtains the h-axis reference currentdifference Δihc using the following formula on the basis of the currentchange per unit voltage.Δihc=(½)·Δt·(1/Ldc+1/Lqc)sin 2(θv−θc)

Next, the real current difference Δih in the direction of the h axis isobtained by the h-axis current differential calculation unit 24.

Step 131

The V-phase and W-phase current differences Δiva and Δiwa in the sectionA are calculated. Symbols iv(ta), iv(tb), and iv(tc) indicate V-phasecurrents respectively at the times ta, tb, and tc shown in FIG. 13. Inthe same way, symbols iw(ta), iw(tb), and iw(tc) indicate W-phasecurrents respectively at the times ta, tb, and tc.

Step 132

The V-phase and W-phase current differences Δivb and Δiwb in the sectionB are calculated.

Step 133

From the difference between the current differences in the sections Aand B, the V-phase and W-phase current differences Δiv and Δiw areobtained. Further, the U-phase current difference Δiu is obtained bycalculation of (−Δiv−Δiw). These current differences indicate changes inthe current due to the voltage difference vector ΔVs which is thedifference in the applied voltages in the sections A and B.

Step 134

On the basis of the 3-phase current differences Δiu, Δiv, and Δiw, theh-axial phase θh, and the absolute value ΔVs0 of the voltage differencevector, the h-axial current difference Δih is obtained. In this case,division by the absolute value ΔVs0 means the current difference perunit voltage.

The difference between the h-axial current difference Δih and theh-axial reference current difference Δihc obtained in this way is thedifference between the rotor position θ and the inferred rotor positionθc, so that the inferred rotor position θc can be converged to the rotorposition θ using the proportion-integration calculation by the positiondetection unit 13 so as to set the difference between the h-axialcurrent difference Δih and the h-axial reference current difference Δihcto 0. This principle is the same as that of the first embodiment.However, the reason that the function block is complicate is that thecurrent change of the rotatory coordinate system of the h axis isdetected instead of the current change in the direction of the α axis(U-phase direction).

The detection principle of the rotor position will be explained indetail by referring to the concrete vector diagram shown in FIG. 17.FIG. 17 shows, in the same way as with FIG. 5, the condition that theinferred rotor position θc inferred by the controller 4 is shifted inthe direction proceeding more than the real rotor position θ. Thevoltage difference vector ΔVs varies with the control voltage vector Vc,so that in FIG. 17, a case that the phase θvc of the control voltagevector Vc is 150 degrees will be explained.

The voltage difference vector ΔVs in this case, as shown in FIG. 14(e),is directed in the direction of −120 degrees. For the voltage differencevector ΔVs, the real current differential vector Δi and the referencecurrent differential vector Δic are respectively the arrow of a solidline and the arrow of a dashed line shown in FIG. 17. Therefore, thedetection current differential vector Δic, as shown in FIG. 17, isdirected in the direction of the first quadrant. Then, in considerationof the h axis obtained from Formula 21, it is found that it is the thirdquadrant close to −90 degrees. Therefore, the h-axial component of thedetection current differential vector Δis is negative and it may be saidthat the real rotor position θ is smaller than the inferred rotorposition θc. When the value is input to the position detection unit 13,the position detection unit 13 calculates so as to make the inferredrotor position θc smaller, so that it gradually approaches the realrotor position θ.

Next, even when the control voltage vector Vc is different from theexample shown in FIG. 17, it will be explained by referring to FIGS. 18and 19 that position detection is possible.

As shown in FIG. 18(e), when the control voltage vector Vc is at 170degrees, the voltage difference vector ΔVs is directed in the directionclose to −150 degrees. Further, FIG. 18(f) shows that the absolute valueΔVs0 thereof is made smaller than that shown in FIG. 17. FIG. 19 is avector diagram at this time.

The control voltage vector Vc and the voltage difference vector ΔVs arechanged compared with the case shown in FIG. 17. Therefore, the currentdifferential vector Δi and the reference current differential vector Δicare also changed. Actually, the absolute value of the voltage differencevector ΔVs is reduced and the magnitudes of the current differentialvector Δi and the reference current differential vector Δic are alsochanged. FIGS. 17 and 19 show current changes per unit voltage. Thisprocess is the one executed at Step 134 shown in FIG. 16.

Due to such a relation, the detection current differential vector Δis isdirected in the direction of the second quadrant. On the other hand, theh axis is in the fourth quadrant, so that the h-axial component of thedetection current differential vector Δis has a negative value in thesame way as with FIG. 17. Therefore, it is found that the inferred rotorposition θc at that time proceeds than the real rotor position θ.Namely, when the detection current difference component in the directionof the h axis is detected regardless of the direction of the controlvoltage vector Vc, the shift of the rotor position can be inferred. Bythis method, the shift of the rotor position can be detected everyperiod of the PWM signal using the saliency of the rotor, so that therotor position can be inferred at high speed.

According to the fourth embodiment, from the waveform of the PWM signalgenerated by the control voltage, a voltage difference equivalent to thedetection voltage can be obtained, so that the current changingcondition for the voltage difference is detected without applying thedetection voltage and the rotor position can be inferred at high speed.Therefore, according to the fourth embodiment, a rapid-response positionsensorless control system can be realized free of noise generated byaddition of the detection voltage and an increase in loss.

The fifth embodiment of the present invention will be explained byreferring to FIG. 20. The fifth embodiment has a constitution ofchanging the calculation method for detection of the rotor positionaccording to the motor speed ω. The fourth embodiment shown in FIG. 12can detect the rotor position without applying the detection voltage.However, the voltage difference vector equivalent to the detectionvoltage varies with the magnitude of the control voltage, so that in thelow torque operation state at low speed, the voltage difference vectorreduces and the position detection precision lowers. On the other hand,the detection methods shown in FIG. 1 (the first embodiment), FIG. 8(the second embodiment), and FIG. 11 (the third embodiment) are a methodfor applying the detection voltage vector, so that they can detect theposition precisely even in the stop state and low speed state.Therefore, the fifth embodiment has a constitution that the detectionmethod is switched according to the motor speed ω, thus convenientposition detection is realized.

A main difference of the fifth embodiment from the fourth embodimentshown in FIG. 12 is that a mode decision unit 30 is installed, and thecalculation contents of the voltage setting unit 25 are changedaccording to the operation mode, and current detection is executed bythe U-phase current sensor 5 u and the V-phase current sensor 5 v.Changing of the current sensor is made for simple explanation of thefifth embodiment and the phase of current detection is not limited.

The function of the mode decision unit 30 will be explained by referringto the flow chart shown in FIG. 21.

Step 141

The mode decision unit 30 inputs the motor speed ω from the speeddetection unit 14.

Step 142

The mode decision unit 30 compares the absolute value of the motor speedω with the first speed ω1 and branches the process.

Step 143

When the absolute value of the motor speed ω is not lower than the firstspeed ω1, the mode decision unit 30 compares the absolute value with thesecond speed ω2 and branches the process.

Step 144

When the absolute value of the motor speed ω is lower than the firstspeed ω1, the mode decision unit 30 sets the mode MD to 1 (means thatthe synchronous motor 1 is in the low speed state including stop).

Step 145

When the absolute value of the motor speed ω is lower than the secondspeed ω2, the mode decision unit 30 sets the mode MD to 2 (means thatthe synchronous motor 1 is in the intermediate speed state).

Step 146

When the absolute value of the motor speed ω is the second speed ω2 orhigher, the mode decision unit 30 sets the mode MD=3 meaning the highspeed state.

The PWM signal generation unit 8 and the voltage setting unit 25 inputthe set mode MD.

The PWM signal generation unit 8, when the mode MD=1, outputs thecarrier synchronizing signal P1 for setting the timing of currentdetection and when the mode MD=2 or MD=3, the PWM signal generation unit8 outputs the carrier synchronizing signal P2. The relation between thecarrier synchronizing signals P1 and P2 and the carrier is as explainedby referring to FIGS. 2 and 13. This means that the mode MD=1 is basedon the method of the first embodiment shown in FIG. 1 and the mode MD=2and MD=3 are based on the method of the fourth embodiment shown in FIG.12.

The processing function of the voltage setting unit 25 will be explainedby referring to FIG. 22. A difference from the voltage setting unit 25shown in FIG. 15 is that a plurality of voltage calculation units and aswitching unit are provided. The switching unit 37 selects and outputscalculation results of the first voltage calculation unit 35, the secondvoltage calculation units 33, and the third voltage calculation unit 31according to the mode MD=1, 2, or 3.

When the speed ω of the synchronous motor 1 is in the low mode MD=1, thevoltage setting unit 25 selects the first voltage calculation unit 35and executes the calculation close to the process of the secondembodiment shown in FIG. 8. The voltage setting unit 25 outputs the sinewave voltages shown in Fig. (a) for the voltage phase θvc, multipliesthem by the absolute value Vc0 of the control voltage vector by themultiplication unit 36, thereby sets the unit voltage of each phaseproportional to the control voltage to the control voltages of eachphase Vuc, Vvc, and Vwc. Further, the voltage setting unit 25 decidesthe detection voltages of each phase Vus, Vvs, and Vws by the inferredrotor position θc by the method shown in FIG. 3 and processes so as tooutput them from the PWM signal generation unit 8 at the timing of thecarrier synchronizing signal P1. The absolute value Vs1 of the detectionvoltage equivalent to the absolute value ΔVs0 of the voltage differencevector is constant, so that this value is output as the absolute valueΔVs0 of the voltage difference vector. Further, the detection voltagedirection θv equivalent to the phase of the voltage difference vector isoutput by calculating (2θc+π/2). As Formula 21 shows, this means thatthe h-axial direction θh is 0, that is, it is the U-phase direction.Therefore, performing of the process of the first voltage calculationunit 35 is performing of the same calculation as that of the secondembodiment shown in FIG. 8.

When the speed ω of the synchronous motor 1 is in the middle speed modeMD=2, the voltage setting unit 25 selects the second voltage calculationunit 33 and executes the calculation. The calculation basically executesthe same process as that of the voltage setting unit 25 of the fourthembodiment shown in FIG. 15, though it is a difference that the tableused for calculation is prepared on the basis of FIG. 24(a) and (f).

In FIG. 24, the voltage of each phase has a waveform that a 0-phasevoltage of triple harmonic wave is added to the normal sine wavevoltage. The reason of adoption of the waveform is that as shown in FIG.24(f), for the phase θvc of the control voltage vector, the absolutevalue of the voltage difference vector ΔVs is changed little and thephase is changed almost constantly. Therefore, the position detectionprecision can be ensured stably and the control in the middle speedrotation region can be stabilized. Further, the voltages as shown inFIG. 24(a) are ones with the 0-phase voltage added, so that the currentwaveform of each phase will not be adversely affected.

When the speed ω of the synchronous motor 1 further increases andreaches the high-speed rotation state mode MD=3, the voltage settingunit 25 selects the calculation by the third voltage calculation unit31. In this case, the table for performing the voltage calculation ofeach phase is prepared on the basis of FIG. 23(a) and (f). As FIG. 23shows, when the phase of the 0-phase voltage of the third harmonic waveis reversed 180 degrees, the maximum value of the voltage of each phaseis reduced. By doing this, when the synchronous motor 1 rotates at highspeed and the counter electromotive force increases, the voltage userate of the inverter 3 can be improved. As shown in FIG. 24(f), there isa disadvantage that the absolute value of the voltage difference vectorvaries greatly, while there is an advantage that the voltage range forrealizing stable control can be enlarged.

The fifth embodiment can precisely detect the rotor position within thewide range from the stop state to the high-speed rotation state of thesynchronous motor 1 and execute rapid-response control.

Further, it is basically desirable for this method to together usepolarity discrimination for judging the N pole or S pole.

With respect to the synchronous motor 1 of each embodiment mentionedabove, the motor including the rotor having reverse saliency isexplained. However, also to a synchronous motor or a reluctance motorhaving saliency, the present invention can be applied using thesaliency. Further, also to an induction motor, the present invention canbe applied by making the reluctance different between the magnetic fluxdirection and the perpendicular direction to it from the magneticsaturation characteristic by the magnetic flux.

Further, needless to say, in consideration of the effect by rotation ofthe rotor of the motor during the sampling time, the pole position maybe calculated.

Pole position detection is not limited to the method for executing everyone period or two periods of the carrier and a method for detecting thepole position every multi-period of the carrier using current changesand a method for detecting the pole position on the basis of currentchanges in units of a plurality of periods can be executed.

This control not only can be applied to a driving AC motor for anelectric car or a hybrid car but also can be widely applied as aposition sensorless control system of an AC motor.

The present invention can realize a rapid-response motor controllerwithout using a pole position sensor for detecting the rotation positionof a rotor.

1. A motor controller comprising an AC motor, a power converter forapplying a voltage to said AC motor by a PWM signal generated bycomparing a command value with a carrier, and a controller for detectinga rotor position of said AC motor and controlling said command value,wherein: said controller detects said rotor position on the basis of adifference between a real current differential vector and a referencecurrent differential vector.
 2. A motor controller comprising an ACmotor, a power converter for applying a voltage to said AC motor by aPWM signal generated by comparing a voltage command value with acarrier, and a controller for detecting a rotor position of said ACmotor and controlling said voltage command value, wherein: saidcontroller has a first phase current detection unit for detectingrespective current differentials changed by respective voltage vectorsapplied in a plurality of sections, a reference phase currentdifferential calculation unit for calculating reference phase currentdifferentials obtained by a plurality of voltage vector differences, anda position detection unit for detecting said rotor position of said ACmotor using said current differentials and said reference phase currentdifferentials.
 3. A motor controller according to claim 2, wherein saidfirst phase current detection unit has a function for removing a DCcomponent.
 4. A motor controller according to claim 2, wherein saidcontroller has a second phase current detection unit for detecting phasecurrents of two phases unlike said first phase current detection unit.5. A motor controller according to claim 4, wherein said controller hasan error detection unit for detecting an error in said phase currentdetection units on the basis of AC components of said phase currentsdetected by said first and second phase current detection units.
 6. Amotor controller comprising an AC motor, a power converter for applyinga voltage to said AC motor by a PWM signal generated by comparing avoltage command value with a carrier, and a controller for detecting arotor position of said AC motor and controlling said voltage commandvalue, wherein: said controller has a current detection unit fordetecting respective current differential vectors changed by respectivevoltage vectors applied in a plurality of sections, a reference currentdifferential vector calculation unit for calculating reference currentdifferential vectors obtained by said plurality of voltage vectordifferences, and a position detection unit for detecting said rotorposition of said AC motor using differences between said currentdifferential vectors and said reference current differential vectors. 7.A motor controller according to claim 6, wherein said current detectionunit detects currents at a plurality of points of time when said carrierreaches close to a center value and obtains said current differentialvectors on the basis of said plurality of currents.
 8. A motorcontroller according to claim 7, wherein said controller has a detectionphase changing unit for changing a detection phase by adding azero-phase voltage to said voltage.
 9. A motor controller according toclaim 6, wherein in said controller, a detection phase decision unit forcalculating a detection phase decided by said voltage and said detectedrotor position has a position detection unit for detecting said rotorposition using detection phase components of said differences of saidcurrent differential vectors and detection phase components of saidreference current differential vectors.
 10. A motor controller accordingto claim 9, wherein said controller has a detection phase changing unitfor changing said detection phase by adding a zero-phase voltage to saidvoltage.
 11. A motor controller according to claim 6, wherein saidcontroller switches current detection timing according to a motor speed.12. A motor controller comprising an AC motor, a power converter forapplying a voltage to said AC motor by a PWM signal generated bycomparing a voltage command value with a carrier, and a controller fordetecting a rotor position of said AC motor and controlling said voltagecommand value, wherein: said controller has a current detection unit fordetecting a current of said AC motor at a point of time when saidcarrier reaches close to a center value and a position detection unitfor detecting said rotor position of said AC motor using said detectedcurrent.
 13. A motor controller according to claim 12, wherein saidcontroller has a detection phase decision unit for calculating adetection phase decided by said voltage and said detected rotor positionand a position detection unit for detecting said rotor position using adetection phase component of a current vector.
 14. A motor controlleraccording to claim 13, wherein said controller has a detection phasechanging unit for changing said detection phase by adding a zero-phasevoltage to said voltage.
 15. A motor controller comprising an AC motor,a power converter for applying a voltage to said AC motor by a PWMsignal generated by comparing a voltage command value with a carrier,and a controller for detecting a rotor position of said AC motor andcontrolling said voltage command value, wherein: said controller has acurrent detection unit for detecting a current of said AC motor insynchronization with said carrier, a detection phase decision unit forcalculating a detection position decided by said voltage and saiddetected rotor position, and a position detection unit for detectingsaid rotor position by using a component of said detection phase of acurrent vector; and wherein said controller has a detection phasechanging unit for changing said detection phase by adding a zero-phasevoltage to said voltage.
 16. A motor controller comprising an AC motor,a power converter for applying a voltage to said AC motor by a PWMsignal generated by comparing a voltage command value with a carrier,and a controller for detecting a rotor position of said AC motor andcontrolling said voltage command value, wherein: said controller has acurrent detection unit for detecting a current of said AC motor insynchronization with said carrier, a detection phase decision unit forcalculating a detection position decided by said voltage and saiddetected rotor position, and a position detection unit for detectingsaid rotor position by using a component of said detection phase of acurrent vector; and wherein said controller switches current detectiontiming according to a motor speed.
 17. A motor controller according toclaim 16, wherein said current detection timing is a point of time whensaid carrier reaches a maximum value or a minimum value when said motorspeed is low and a point of time when said carrier reaches close to acenter value when said motor speed is high.
 18. A motor controllercomprising an AC motor, a power converter for applying a voltage to saidAC motor by a PWM signal generated by comparing a command value with acarrier, and a controller for detecting a rotor position of said ACmotor and controlling said command value, wherein: said controller has acalculation unit for deciding a detection voltage to be added to saidcommand value on the basis of said detected rotor position and decidinga reference current variation which is a standard for current changes, afirst phase current detection unit for detecting a current variation forsaid detection voltage, and a position detection unit for detecting saidrotor position of said AC motor on the basis of said reference currentvariation and said current variation.
 19. A motor controller comprisingan AC motor, a power converter for applying a voltage to said AC motorby a PWM signal generated by comparing a voltage command value with acarrier, and a controller for detecting a rotor position of said ACmotor and controlling said voltage command value, wherein: saidcontroller has a current detection unit for detecting a current of saidAC motor in synchronization with said carrier, a detection phasedecision unit for calculating a detection position decided by saidvoltage and said detected rotor position, and a position detection unitfor detecting said rotor position by using a component of said detectionphase of a current vector; wherein said current detection unit detects acurrent at a point of time when said carrier reaches close to a centervalue; and wherein said controller has a detection phase changing unitfor changing said detection phase by adding a zero-phase voltage to saidvoltage.